Systems and methods of rf power transmission, modulation, and amplification, including control functions to transition an output of a miso device

ABSTRACT

Embodiments of the present invention include a method and system for control of a multiple-input-single output (MISO) device. For example, the method includes partitioning a waveform constellation space into a plurality of regions, where each region of the plurality of regions is associated with one or more control functions of the MISO device. The method also includes transitioning the MISO device between a plurality of classes of operation based on the one or more control functions.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a continuation of U.S. application Ser. No.13/565,007, filed Aug. 2, 2012, now allowed, which is a continuation ofU.S. application Ser. No. 13/069,155, filed Mar. 22, 2011, now U.S. Pat.No. 8,410,849, which is a continuation of U.S. patent application Ser.No. 12/236,079, filed Sep. 23, 2008, now U.S. Pat. No. 7,911,272, whichis a continuation-in-part of U.S. patent application Ser. No.12/142,521, filed Jun. 19, 2008, now U.S. Pat. No. 8,013,675, whichclaims the benefit of U.S. Provisional Patent Application No.60/929,239, filed Jun. 19, 2007, and U.S. Provisional Patent ApplicationNo. 60/929,584, filed Jul. 3, 2007, all of which are incorporated hereinby reference in their entireties.

The present application is related to U.S. patent application Ser. No.11/256,172, filed Oct. 24, 2005, now U.S. Pat. No. 7,184,723 and U.S.patent application Ser. No. 11/508,989, filed Aug. 24, 2006, now U.S.Pat. No. 7,355,470, both of which are incorporated herein by referencein their entireties.

BACKGROUND

1. Field

Embodiments of the present invention relate generally to RF (radiofrequency) power transmission, modulation, and amplification.

2. Background

Today's RF power amplifiers are required to generate complex RF signalswith stringent output power and linearity requirements. For example, inorder to comply with the requirements of a WCDMA waveform, a poweramplifier needs to support approximately 30-40 dB of instantaneousoutput power dynamic range at a given power output level. This is mainlydue to the ACPR (Adjacent Channel Power Ratio) and the ACLR (AdjacentChannel Leakage Ratio) requirements of the WCDMA waveform, which requirevery deep nulls as the output power waveform crosses zero.

Generally, the ACLR and ACPR that a power amplifier can achieve arerelated to the linearity of the power amplifier over the output powerrange of the desired waveform. Modern RF waveforms (e.g., OFDM, CDMA,WCDMA, etc.) are characterized by their associated PAP (Peak-to-AveragePower) ratios. As such, in order to generate such waveforms, the poweramplifier needs to be able to operate in a largely linear manner over awide output power range that encompasses the output power range of thedesired waveforms.

Outphasing amplification or LINC (Linear Amplification with NonlinearComponents) provides an amplification technique with the desirablelinearity to amplify RF waveforms with large PAP ratios. Outphasingworks by separating a signal into equal and constant envelopeconstituents, linearly amplifying the constituents, and combining theamplified constituents to generate the desired output signal. Topreserve linearity when combining the amplified constituents, existingoutphasing techniques use an isolating and/or a combining element, whichprovides the needed isolation between the branches of the outphasingamplifier to reduce non-linear distortion.

In several respects, however, existing outphasing techniques are notsuitable for implementation in modern portable devices. For example, theisolating and/or combining element that they use causes a degradation inoutput signal power (due to insertion loss and limited bandwidth) and,correspondingly, low power amplifier efficiency. Further, the typicallylarge size of isolating/combining elements precludes having them inmonolithic amplifier designs.

There is a need therefore for outphasing amplification systems andmethods that eliminate the isolating/combining element used in existingoutphasing techniques, while providing substantially linearamplification over a wide output power dynamic range to support modernRF waveforms.

BRIEF SUMMARY

Embodiments of the present invention relate generally to RF powertransmission, modulation, and amplification.

An embodiment of the present invention includes a method for control ofa multiple-input-single-output (MISO) device. The method can includepartitioning a waveform constellation space into a plurality of regions,where each region of the plurality of regions is associated with one ormore control functions of a multiple-input-single-output (MISO) device.The method can also include transitioning the MISO device between aplurality of classes of operation based on the one or more controlfunctions.

Another embodiment of the present invention includes a system. Thesystem includes a multiple-input-single-output (MISO) device and atransfer function module. The transfer function module is configured totransition the MISO device between a plurality of classes of operationbased on one or more control functions, wherein the one or more controlfunctions are each associated with a region from a plurality of regionspartitioned from a waveform constellation space.

Further embodiments, features, and advantages of the present invention,as well as the structure and operation of the various embodiments of thepresent invention, are described in detail below with reference to theaccompanying drawings.

BRIEF DESCRIPTION OF THE FIGURES

The accompanying drawings, which are incorporated herein and form a partof the specification, illustrate the present invention and, togetherwith the description, farther serve to explain the principles of theinvention and to enable a person skilled in the pertinent art to makeand use the invention.

FIG. 1 illustrates the theoretical error-free normalized amplitude of aphasor as a function of the differential phase between the constituentsof the phasor.

FIG. 2 illustrates the theoretical error in the amplitude of a phasor asa function of the differential phase between the constituents of thephasor, when the constituents are combined with infinite isolation.

FIG. 3 illustrates the theoretical error in the amplitude of a phasor asa function of the differential phase between the constituents of thephasor, when the constituents are combined with 20 of dB isolation.

FIG. 4 compares the theoretical error-free normalized amplitude of aphasor and the theoretical amplitude of the phasor when the constituentsare combined with 20 of dB isolation, as a function of the differentialphase between the constituents of the phasor.

FIG. 5 compares the theoretical error-free normalized amplitude of aphasor and the theoretical normalized amplitude of the phasor when theconstituents are combined with 20 dB of isolation, as a function of thedifferential phase between the constituents of the phasor.

FIG. 6 compares the theoretical error-free normalized amplitude of aphasor and the theoretical normalized amplitude of the phasor when theconstituents are combined with 25 dB of isolation, as a function of thedifferential phase between the constituents of the phasor.

FIG. 7 illustrates the derivative of the theoretical error in theamplitude of a phasor as a function of the differential phase betweenthe constituents of the phasor, when the constituents are combined with25 dB of isolation.

FIG. 8 illustrates the derivative of the theoretical phasor error in theamplitude of a phasor as a function of the differential phase betweenthe constituents of the phasor, when the constituents are combined with30 dB of isolation.

FIG. 9 compares blended control amplification according to an embodimentof the present invention and outphasing amplification, with respect tothe level of control over the constituents of a desired phasor.

FIG. 10 illustrates an example blended control amplification systemaccording to an embodiment of the present invention.

FIG. 11 illustrates the relationship between the error (in amplitude andphase) in a phasor and the imbalance (in amplitude and phase) betweenthe constituents of the phasor.

FIG. 12 illustrates the power output associated with a phasor as afunction of the differential phase between the constituents of thephasor, when no imbalance in amplitude and phase exists between theconstituents of the phasor.

FIG. 13 compares the power output associated with a phasor as a functionof the differential phase between the constituents of the phasor, fordifferent scenarios of amplitude and phase imbalance between theconstituents of the phasor.

FIG. 14 compares the amplitude error in the power output associated witha phasor as a function of the differential phase between theconstituents of the phasor, for different scenarios of amplitude andphase imbalance between the constituents of the phasor.

FIG. 15 compares the phase error in the power output associated with aphasor as a function of the differential phase between the constituentsof the phasor, for different scenarios of amplitude and phase imbalancebetween the constituents of the phasor.

FIG. 16 illustrates an example blended control amplification functionaccording to an embodiment of the present invention.

FIG. 17 illustrates real-time blended control amplification for anexample output power waveform according to an embodiment of the presentinvention.

FIG. 18 is an example that illustrates the output stage theoreticalefficiency of a blended control amplification system according to anembodiment of the present invention, as function of the output stagecurrent.

FIG. 19 compares the output power transfer characteristic of a blendedcontrol amplification system according to an embodiment of the presentinvention and the output power transfer characteristic of an idealoutphasing amplification system.

FIGS. 20-23 illustrate using a blended control amplification functionaccording to an embodiment of the present invention to generate anexample modulated ramp output.

FIGS. 24-26 illustrate example blended control methods according toembodiments of the present invention.

The present invention will be described with reference to theaccompanying drawings. Generally, the drawing in which an element firstappears is typically indicated by the leftmost digit(s) in thecorresponding reference number.

DETAILED DESCRIPTION

In commonly owned U.S. patent(s) and application(s), cross-referencedabove, VPA (Vector Power Amplification) and MISO(Multiple-Input-Single-Output) amplification embodiments wereintroduced. VPA and MISO provide combiner-less RF power amplification,which results in high power amplifier efficiency. At the same time,despite minimal or zero branch isolation, VPA and MISO amplificationinclude innovative amplifier bias functions that effectively result inhighly linear amplification over the entire output power range ofdesired waveforms.

In the following sections, embodiments of a blended control function foroperating a MISO amplifier embodiment are provided. The blended controlfunction allows for the mixing of various output power control functionsenabled by VPA and MISO, to generate a desired waveform with highaccuracy. In Section 2, the relationship between branch isolation (i.e.,isolation between the branches of an outphasing amplifier) and outputpower error is described. This serves as an introduction to thepractical limitations of a pure outphasing system, which are describedin Section 3. In Section 4, blended control amplification is introduced.In Section 5, design considerations related to blended controlamplification are described. Section 6 describes an example blendedcontrol function and associated performance results. Finally, Section 7presents example blended control methods according to embodiments of thepresent invention.

1. RELATIONSHIP BETWEEN BRANCH ISOLATION AND OUTPUT POWER ERROR

Equation (1) below describes the sum of two sine waves (or phasors) ofequal amplitude, A, and frequency, ω_(c), but having a differentialphase θ:

R sin(ω_(c) t+δ)=A sin ω_(c) t+A sin(ω_(c) t+θ).  (1)

The resulting phasor has amplitude R and phase 8. Equation (1) furtherindicates that any desired phasor of given amplitude and phase can beobtained from the sum of two equal amplitude phasors with appropriatedifferential phase between the two. The equal amplitude phasors arecommonly referred to as the constituents of the desired phasor.

From equation (1), it can be further noted that the amplitude of theresulting phasor is a function of the differential phase, θ, between itsconstituents, as follows:

$\begin{matrix}{{R(\theta)} = {A{\frac{{\sin \left( {\omega_{c}t} \right)} + {\sin \left( {{\omega_{c}t} + \theta} \right)}}{\sin \left( {{\omega_{c}t} + {\delta (\theta)}} \right)}.}}} & (2)\end{matrix}$

Similarly, the phase, δ(θ), of the resulting phasor is a function of thedifferential phase, θ, between its constituents.

FIG. 1 illustrates the theoretical error-free normalized amplitude of aphasor as a function of the differential phase between its constituents.As shown, the differential phase is swept from 0 degrees toapproximately 150 degrees. At zero degrees, the constituents arephase-aligned with each other and result in a maximum normalized phasoramplitude of 2. At approximately 150 degrees, the constituents areseparated in phase by approximately 150 degrees and result in anormalized phasor amplitude of approximately 0.5.

In the context of power amplification, phasor amplitude curve 102 shownin FIG. 1 may represent the output power amplitude of an outphasingpower amplifier as a function of the differential phase between theconstituents of the output waveform. Thus, the output power dynamicrange spanned by the amplitude curve 102 would be approximately 12 dB(20 log (0.5/2)). More particularly, phasor amplitude curve 102 wouldrepresent the output power amplitude generated by an ideal outphasingpower amplifier. In other words, phasor amplitude curve 102 would resultwhen infinite isolation and infinite vector accuracy exists between thebranches of the outphasing amplifier. As described above, however, thisis impractical to design due to the power, cost, and size inefficienciesintroduced by isolating/combining elements, which are typically used inconventional outphasing systems. Alternatively, if noisolating/combining element is used, the branches of the outphasingamplifier would have to be located on distinct substrates, whichnecessarily precludes a monolithic, compact design suitable for today'sportable devices.

Therefore, for practical outphasing amplifier designs, a finiteisolation between the branches of an outphasing amplifier is to beassumed. This finite isolation results in crosstalk between the branchesof the amplifier (i.e., the signal in one branch causes an undesiredeffect on the signal in the other branch), effectively causing an errorsignal to appear at the output of the power amplifier.

In the worst case scenario, the crosstalk between the branches of theamplifier is entirely non-linear. The resulting error signal at theoutput of the power amplifier can therefore be written as:

R _(nonlinear) sin(ω_(c) t+δ)=A _(a) sin(ω_(c) t)·A _(b) sin(ω_(c)t+θ))+A _(a) sin(ω_(c) t+θ)·A _(b) sin(ω_(c) t)  (3)

where A_(a) represents the desired phase amplitude and A_(b) representsthe amplitude of the crosstalk between the branches of the amplifier.

A_(a) and A_(b) are related to each other according to A_(b)=1−A_(a)(where the sum of A_(a) and A_(b) is normalized to 1). The relativeisolation in dB between the branches of the amplifier can be calculatedas −20 log (A_(b)). For example, for A_(a)=0.5, A_(b)=0.5 and therelative isolation is −20 log (0.5)=6 dB.

From equation (3) above, the amplitude of the error signal at the outputof the amplifier is a function of the differential phase, θ, and can bedescribed as:

$\begin{matrix}{{R_{nonlinear}(\theta)} = {2\; A_{a}{{\sin \left( {\omega_{c}t} \right)} \cdot A_{b}}{\frac{\sin \left( {{\omega_{c}t} + \theta} \right)}{\sin \left( {{\omega_{c}t} + {\delta (\theta)}} \right)}.}}} & (4)\end{matrix}$

In equation (4), A_(a)=1 or equivalently A_(b)=0 corresponds to infiniteisolation between the branches of the amplifier. The error amplitudewould thus be zero as illustrated in FIG. 2, which shows the erroramplitude as a function of the differential phase, θ, between theconstituents of the desired phasor.

For A_(a)=0.9 or equivalently A_(b)=0.1, the branch isolation is 20 dB(−20 log (0.1)) and the error amplitude as a function of θ is asdescribed by error amplitude curve 302 in FIG. 3. As shown in FIG. 3,the error amplitude is near zero for small values of θ but begins todeviate from zero as θ moves away from zero. This is because as θincreases, the amplitude of the desired phasor decreases and the effectof crosstalk between the branches becomes greater.

2. PRACTICAL LIMITATIONS OF PURE OUTPHASING

From the resulting error amplitude curve 302 of FIG. 3, the amplitude ofthe phasor when the constituents are combined with 20 dB of isolationcan be determined. This is illustrated in FIG. 4, which compares thetheoretical error-free normalized amplitude of a phasor (curve 102) andthe theoretical amplitude of the phasor when the constituents arecombined with 20 of dB isolation (curve 402), as a function of thedifferential phase between the constituents of the phasor. Amplitudecurve 402 is obtained by summing error amplitude curve 302 anderror-free normalized amplitude curve 102.

As shown in FIG. 4, curve 402 deviates from curve 102 over most of thedifferential phase range, illustrating the effect of the non linearcrosstalk error on the amplitude of the desired phasor. Note, however,that curve 402 is not normalized as curve 102.

In FIG. 5, curve 402 is normalized such that the maximum amplitudecorresponds to the value of 2, resulting in normalized amplitude curve502. As shown in FIG. 5, curve 502 and curve 102 align with one anotherover a portion of the differential phase range (approximately 50 degreesin each direction moving away from 0 degrees). This indicates that inthe worst case scenario (i.e., error entirely nonlinear), even with only20 dB of branch isolation, a pure outphasing system (i.e., system thatrelies exclusively on modulating the phases of the constituent phasorswith no additional calibration) matches the performance of an idealoutphasing system with infinite isolation (i.e., provide comparablelinear amplification) over a portion of the differential phase range.From an output power range perspective, the pure outphasing systemmatches the waveform performance of an ideal outphasing system over aportion of the output power dynamic range of the desired waveform. InFIG. 5, this is approximately 2.5 dB of output power control range (−20log (1.5/2)).

Increasing the branch isolation to 25 dB would further increase theoutput power control range that can be achieved using only a pureoutphasing system. This is shown in FIG. 6, which compares the desiredphasor normalized amplitude with 25 dB of branch isolation (curve 602)and the theoretical error-free normalized amplitude (curve 102). Curve604 is the error amplitude as function of the differential phase with 25dB of branch isolation. As shown in FIG. 6, curves 102 and 602 arealigned with each other over an even larger portion of the differentialphase range. From an output power range perspective, this isapproximately 6 dB of output power control range, over which the pureoutphasing system (with 25 dB of branch isolation) and an idealoutphasing system (infinite isolation) would achieve identicalamplification performance.

The differential phase range over which a pure outphasing system can beused exclusively (while matching the performance of an ideal outphasingsystem) can be further determined by examining the derivative of theerror amplitude as a function of the differential phase. This isillustrated for 25 dB and 30 dB of branch isolation respectively inFIGS. 7 and 8. As shown in FIG. 7, the error amplitude derivative curveis relatively flat between −120 degrees and +120 degrees, whichindicates an insignificant variation in error amplitude over that phaserange. Similarly, with 30 dB of branch isolation, the error amplitudederivative curve is flat over an even larger range of the differentialphase, as shown in FIG. 8.

It should be noted that the analysis above represents a worst casescenario because it assumes that the crosstalk error is entirelynonlinear. In practice, a portion of the crosstalk error will be linear,which further increases the differential phase range over which pureoutphasing can be used with no additional calibration or, alternatively,allows for lower branch isolation to be used. What can be further notedis that a pure outphasing system can be used to generate a portion ofthe output power range of a desired waveform with comparable performanceto an ideal outphasing system. For waveforms with small output powerdynamic range, pure outphasing may be used exclusively to generate suchwaveforms. However, for waveforms with larger output power dynamicrange, practical limitations (i.e., finite branch isolation, crosstalk,etc.) may preclude the use of a pure outphasing solution when highlyaccurate, distortion-free amplification is desired.

3. BLENDED CONTROL AMPLIFICATION

In this section, a blended control amplification approach according toan embodiment of the present invention will be presented. The blendedapproach combines pure outphasing with bias and/or amplitude control toyield an accurate, practical, and producible system with substantiallycomparable performance to that of an ideal outphasing system, butwithout the extreme isolation and accuracy requirements of outphasingalone. The blended approach provides a high degree of control over theconstituent phasors (whether in terms of amplitude and/or phase) inorder to generate the desired phasor. This allows for a reduction inboth the branch isolation requirements and the phase/amplitude accuracyrequirements (as related to the constituent phasors) as compared to apure outphasing or ideal outphasing system.

A comparison between the blended approach of the present invention andpure outphasing with respect to the level of control over constituentphasors is provided in FIG. 9.

As shown in FIG. 9, using pure outphasing, the constituent phasors arerestricted in amplitude in that they must fall on the unit circle. Inother words, the only controllable parameter in generating a desiredphasor is the differential phase between the constituent phasors. As aresult, in order to accurately generate a desired waveform, highaccuracy in terms of the differential phase is needed. However, asdescribed above, when the objective is to reduce branch isolation andgenerate complex waveforms with large PAP ratios, accuracy requirementsbecome very stringent as to become almost impractical. This isespecially the case when generating a waveform with a deep null (e.g.,30-40 dB null), which requires the constituents to be exactly phasedifferenced by 180 degrees (i.e., differential phase is 180 degrees) andat which point the error amplitude is greatest, as can be noted fromFIGS. 5-8, for example.

On the other hand, using the blended approach of the present invention,the constituent phasors can be varied both in terms of phase andamplitude to generate the desired waveform. As a result, not only canany desired phasor be generated without having the differential phaseexceed a given amount (e.g., limiting the differential phase to therange over which the error is negligible), but also the amplitude of theconstituent phasors can be reduced at given output levels, whichincreases the operational output power range and repeatability of theoverall system.

In FIG. 9, an example control range of the constituent phasors accordingto an embodiment of the present invention is provided by the shadedcircle area contained within the unit semi-circle. As shown, when thedesired phasor amplitude is large (i.e., high output power), theamplitude of the constituent phasors approaches the radius size of theunit circle. In other words, for large output power levels, theamplitude of the constituent phasors under the blended control approachis comparable to its corresponding amplitude under a pure outphasingapproach. However, as the desired phasor amplitude decreases, theamplitude of the constituent phasors recedes from the unit circle andbegins to deviate from its corresponding amplitude under a pureoutphasing approach.

As a result of the blended approach of the present invention, theaccuracy requirements in terms of phase/amplitude of the constituentphasors can be significantly reduced, which accommodates the branchisolations, vector accuracy, and phase accuracy that can be practicallyexpected. For example, in an embodiment of the blended approach of thepresent invention, when the desired output power tends to zero, theconstituent phasors are also driven to zero amplitude, which essentiallyeliminates any accuracy requirements regarding the differentialamplitude and phase between the constituent phasors or, in other words,entirely reduces the system's sensitivity to branch phase imbalance, forthat particular output power range.

Another advantage of the blended approach of the present invention canalso be gleaned from FIG. 9. This relates to the ability of the blendedapproach of the present invention to generate any desired phasoramplitude (except for the maximum amplitude) using any one of aninfinite number of Constituent phasor configurations. This is verysignificant when compared to an ideal outphasing system, in which thereexists a Single configuration of the constituent phasors for any desiredphasor amplitude (i.e., the constituent phasors must fall on the unitcircle and are symmetrically opposed to each other relative to thecosine axis).

According to an embodiment of the present invention, the shaping of theconstituent phasors in phase and/or amplitude, as described above, isperformed substantially instantaneously or in real time in accordancewith the desired waveform output power trajectory. In an embodiment,this is performed using a combination of phase, bias, and amplitudecontrols, with the control combination (or blend) dynamically changingaccording to the desired waveform output power trajectory. An exampleamplification system according to an embodiment of the presentinvention, which may be used to implement a blended control approach asdescribed above, is now presented with reference to FIG. 10.

Referring to FIG. 10, example amplification system 1000 uses a MISOamplifier. Amplification system 1000 includes a transfer function 1006,vector modulators 1008 and 1010, driver amplifiers 1014 and 1016, and aMISO amplifier 1018. Further detail regarding embodiments of thesecomponents as well as the operation of system 1000 (according to variousembodiments) can be found in commonly owned related patents andapplications, indicated above in the cross-reference section of thispatent application, and incorporated herein by reference in theirentireties. In addition, the MISO amplifier could be replaced with atraditional Outphasing or LINC output amplifier arrangement whichincludes two power amplifiers and a power combiner.

According to an embodiment, which shall now be described, system 1000includes a blended control implementation, which is implemented as acombination of phase, bias, and amplitude controls. For example, phasecontrol (i.e., control of the phases of the constituent phasors) insystem 1000 can be performed using one or more of transfer functionmodule 1006 and vector modulators 1008 and 1010. Bias control, whichincludes biasing power amplifiers 1620 and 1622 within MISO amplifier1018 to affect the amplitude of the desired phasor, is done via biascontrol signal 1024 generated by transfer function module 1006. Notealso that bias control can be affected at drivers 1014 and 1016 viadriver bias control signal 1026. Amplitude control, which includescontrolling the input signals into MISO amplifier 1018 in order toaffect the amplitude of the constituent phasors, can be performed usingone or more of transfer function module 1006 and drivers 1014 and 1016,for example.

According to embodiments of the present invention, system 1000 may useone or more of phase, bias, and amplitude control with varying degreesof weight given to each type of control according to the desiredwaveform. Example blended control functions according to the presentinvention are described below in Section 6.

FIG. 19 compares the output power transfer characteristic of system 1000and that of an ideal outphasing amplification system. As shown, theoutput power performance of system 1000 is almost identical to that ofan ideal outphasing system. Yet, as described above, system 1000requires only 20-25 dB of branch isolation, and other embodiments mayrequire less.

4. PRACTICAL DESIGN CONSIDERATIONS

As would be understood by a person skilled in the art based on theteachings herein, the optimum combination of controls as well as thedegrees of weight given to each type of control within an amplificationsystem according to the present invention will depend on both thecharacteristics of the system itself (e.g., branch isolation,phase/amplitude branch imbalance, etc.) and design consideration such asthe desired waveform output power. Therefore, it is important in orderto design a system with such optimum combination and use of controls tounderstand the practical effects of system characteristics on the outputperformance (i.e., accuracy of the output waveform) of the system.

In the following, the effects of phase and amplitude branch imbalance onthe output performance of an example amplification system according tothe present invention are examined. For ease of analysis andillustration, it is assumed that the constituent phasors (A1 and A2) areconstrained to the first and fourth quadrants of the unit circle, andthat they are designed to be of equal amplitude and symmetrical to eachother with respect to the cosine axis, as illustrated in FIG. 11. Notethat in practice the constituent phasors may occur within any quadrantof the unit circle and are not required to be equal and/or symmetricalto each other with respect to the cosine axis. It is further assumedthat the output power is normalized to a maximum of 30 dB.

Note from the assumptions above that if phasors A1 and A2 are indeedequal in amplitude and symmetrical to each other with respect to thecosine axis (i.e., no amplitude/phase imbalance between the branches ofthe amplifier), the resulting phasor will be perfectly aligned with thecosine axis (i.e., zero phase error in the output waveform). The poweroutput associated with such resulting phasor will be as illustrated inFIG. 12, which shows the power output as a function of the differentialphase between the constituent phasors in an ideal outphasing system.

In practice, however, phase/amplitude branch imbalance cannot beentirely reduced to zero for a variety of reasons, including finitebranch isolation for example, and will affect the choice of combinationof controls. In the analysis below, phase/amplitude branch imbalance isintroduced into an example amplification system according to the presentinvention, and the output performance of the system is examined. Theexample amplification system uses phase control only.

In FIG. 13, the power output associated with a phasor as a function ofthe differential phase between the constituents of the phasor isexamined for various scenarios of phase/amplitude branch imbalance.Power output curve 1302 illustrates the power output with 0 dB ofamplitude imbalance and 0 degrees of phase imbalance between thebranches of the amplification system. In other words, curve 1302illustrates the power output of an ideal outphasing system. Power outputcurve 1304 illustrates the power output for 0.5 dB of amplitude branchimbalance and 5 degrees of phase branch imbalance. Power output curve1306 illustrates the power output for 1 dB of amplitude branch imbalanceand 10 degrees of phase branch imbalance.

As can be seen from FIG. 13, power output curves 1304 and 1306 begin todiverge from power output curve 1302 at differential phase values ofapproximately 80 to 100 degrees.

FIGS. 14 and 15 illustrate the power output amplitude error and thepower output phase error, respectively, as a function of thedifferential phase between the constituents of the phasor, for the samephase/amplitude branch imbalance scenarios as in FIG. 13.

The results from FIGS. 13-15 can be used, based on system designcriteria, to determine an operating range (in terms of differentialphase) over which phase control only can be used. For example, systemdesign criteria may require a maximum allowable power output error of0.5 dB and a maximum allowable power output phase error of 5 degrees.Accordingly, for a system with 0.5 dB of amplitude branch imbalance and5 degrees of phase branch imbalance, the phase control only range wouldbe approximately 0 to 110 degrees (lower of 110 and 140 degrees). Phasecontrol only would thus be able to vary the output power by 4.8 dB witha high degree of accuracy. Similarly, for a system with 1 dB ofamplitude branch imbalance and 10 degrees of phase branch imbalance, thephase control only range would be approximately 0 to 70 degrees (lowerof 70 and 100 degrees). Phase control only would thus be able to varythe output power by 1.7 dB with a high degree of accuracy.

Nonetheless, phase control only would not be able on its own to achieveoutput power control ranges of 30-40 dB, as desired for complexwaveforms, without degrading the accuracy of the desired waveform at lowoutput powers. Therefore, one or more additional types of control (e.g.,bias control, amplitude control) may be needed as used in embodiments ofthe present invention to enable a practical, accurate amplifier designfor complex waveforms.

5. EXAMPLE BLENDED CONTROL FUNCTION AND PERFORMANCE RESULTS

An example blended control function according to an embodiment of thepresent invention will now be presented. The example blended controlfunction is designed to optimize the output performance (i.e., poweroutput accuracy) of an amplification system according to an embodimentof the present invention for a QPSK waveform output. The example blendedcontrol function is illustrated in FIG. 16, wherein it is imposed on topof a QPSK constellation in the complex space defined by cos(wt) andsin(wt). The blended control function partitions the QPSK constellationspace into three control regions 1602, 1604, and 1606, as shown in FIG.16.

In an embodiment, the blended control function determines the type ofcontrol or controls used depending on the instantaneous power of thedesired output waveform. For example, as would be understood by a personskilled in the art, a QPSK signal moves from one constellation point toanother to encode information. However, although all four constellationpoints correspond to equal power, the signal does not moveinstantaneously from one constellation point to another and thus willhave to traverse the trajectory connecting the constellation points, asshown in FIG. 16. Accordingly, the signal will traverse at least twocontrol regions of the blended control function as it moves from anyconstellation point to any other. As it does, the types of controlsapplied within the amplification system to generate the output powerwill also vary.

In an embodiment, the example blended control function of FIG. 16 issuch that control region 1602 is a phase control-biased region (i.e.,higher weight is given to phase control compared to bias control andamplitude control). In another embodiment, control region 1602 is aphase control only region. Control region 1604 is a phase control, biascontrol, and amplitude control region. All three types of controls maybe combined with equal or different weights in control region 1604. Inan embodiment, higher weight is given to bias control than phase controland amplitude control in control region 1604. Control region 1606 is abias control and amplitude control region. Bias control and amplitudecontrol may be combined with equal or different weights in controlregion 1606. In an embodiment, control region 1606 is amplitudecontrol-biased, i.e., amplitude control is given higher weight than biascontrol in control region 1606.

In an embodiment, the example blended control function of FIG. 16enables a variable weighted combination of controls, whereby weightsgiven to each type of control vary according to the desired waveformoutput power. In an embodiment, the variable weighted combination ofcontrols varies from a phase control-biased combination to abias/amplitude control-biased combination as the desired waveform outputpower varies from high to low power levels.

As would be understood by persons skilled in the art, control regions1602, 1604, and 1606 in FIG. 16 are provided for purposes ofillustration only and are not limiting. Other control regions can bedefined according to embodiments of the present invention. Typically,but not exclusively, the boundaries of the control regions are based onthe Complementary Cumulative Density Function (CCDF) of the desiredoutput waveform and the sideband performance criteria. Accordingly, thecontrol regions' boundaries as well as the type of controls used withineach control region can vary according to the desired output waveform,according to embodiments of the present invention.

FIG. 17 illustrates an example output power waveform and a correspondingoutput stage current generated by a MISO amplifier operating accordingto the example blended control function described above. The blendedcontrol function is also shown in FIG. 17 to illustrate, by directmapping, the control region used to generate any given value of theoutput power waveform or the output stage current. For example, when theoutput power waveform goes through a zero crossing, the blended controlfunction is operating in control region 1606.

As shown in FIG. 17, the output stage current closely follows the outputpower waveform. In particular, it is noted that the output stage currentgoes completely to zero when the output power waveform undergoes a zerocrossing. In an embodiment, this corresponds to the MISO amplifier beingoperated in amplitude control-biased control region 1606. In otherwords, the MISO amplifier current is driven to zero by mainlycontrolling the amplitudes of the input, signals of the MISO amplifier.

FIG. 17 further illustrates the MISO amplifier classes of operation as afunction of the output power waveform and the blended control function.As shown, the MISO amplifier transitions between various classes ofoperation (e.g., class S through class A) as the combination of controlsused within the MISO amplifier is varied. For example, the MISOamplifier operates as a class A or B amplifier when the blended controlfunction operates in control region 1606. On the other hand, the MISOamplifier operates in switching mode (class S) when the blended controlfunction operates in control region 1602. This allows for optimizing theefficiency of the MISO amplifier as a function of the instantaneousoutput power of the desired waveform.

FIG. 18 is an example that illustrates the output stage theoreticalpower efficiency as a function of the output stage current for a MISOamplifier operating according to the example blended control functiondescribed above. As shown, the MISO amplifier operates at 100%theoretical efficiency at all times that it operates as a class S-classC amplifier. The MISO amplifier operates at 50% theoretical efficiencywhen it operates as a class A or B amplifier. However, as shown in FIG.18, the MISO amplifier spends very short time operating as class A orclass B amplifier. Accordingly, in an embodiment, the MISO amplifieroperates at 100% theoretical efficiency for 98% (or greater) of the timewhile generating typical cell phone waveforms.

FIGS. 20-23 illustrate using a blended control function to generate anexample modulated ramp output according to an embodiment of the presentinvention. For example, the blended control function may be used withinamplification system 1000 described above.

FIG. 20 illustrates an exemplary desired output amplitude response Asshown, the desired output amplitude transitions linearly from a maximumvalue of 2 to a minimum of zero, before returning linearly to themaximum of 2.

FIG. 21 compares the blended control function and pure outphasing withrespect to the differential phase between the constituent phasors, togenerate the desired output amplitude of FIG. 20. Pure outphasing isrepresented by curve 2102, and the blended control function isrepresented by curve 2104. As shown, for pure outphasing, thedifferential phase spans the entire 180 degrees range, varying from 0degrees to generate the maximum amplitude of 2 to 180 degrees togenerate the minimum amplitude of zero. On the other hand, for theblended control function, the differential phase is restricted to a muchsmaller range (0 to approximately 70 degrees), while other type ofcontrols are also used to generate the desired output. In an embodiment,bias control is used to complement phase control to generate the desiredoutput. Accordingly, the MISO amplifier and/or driver amplifiers thatprecede the MISO amplifier are bias controlled. FIG. 22 illustratesexample bias control signals (represented as voltages 2202 and 2204)provided to bias the MISO amplifier and the driver amplifiers toimplement bias control. For example, voltages 2202 and 2204 may beprovided through bias control signals 1024 and 1026 in amplificationsystem 1000 described above.

Note from FIGS. 21 and 22 that bias control is used at the same time asphase control within the phase control range (0 to 70 degrees), thoughphase control may be used with much higher weight than bias controlwithin that range. This can be noted from voltages 2202 and 2204, whichare modified within the phase control range. Voltages 2202 and 2204continue to vary outside the phase control range and tend to zero as thedesired output amplitude tends to the minimum value of zero. It is notedthat the weights shown in the figures and discussed herein are providedsolely for illustrative purpose and are not limiting. Other weightvalues can be used depending on the situation and the desired outcome.

In an embodiment, when bias control is applied, variations occur in theS (reverse isolation) parameters of the amplifiers of the system,resulting in an associated phase error at the output. Fortunately, thiscan be easily compensated for by applying a rotational transform at thevector modulators of the system. FIG. 22 illustrates the phasemodification applied to compensate for the phase error resulting frombias control. As shown, minimal correction is needed for the first 30 or40 degrees of the differential phase range. This is because bias controlis used with much lower weight than phase control. However, as thedesired output amplitude approaches zero, bias control is used moreheavily and the associated phase error correction becomes greater. Notethat the phase error correction inverts 180 degrees at the zeroamplitude crossing, since the desired output is a single sidebandsuppressed carrier waveform.

6. EXAMPLE BLENDED CONTROL METHODS

FIGS. 24-26 illustrate example blended control methods according toembodiments of the present invention.

FIG. 24 illustrates a process flowchart 2400 of a method for control ina power amplifier. Process 2400 begins in step 2402, which includesdetermining an instantaneous power level of a desired output waveform ofthe power amplifier. In an embodiment, referring to FIG. 10, step 2402can be performed by transfer function module 1006 based on received Iand Q data reflecting the desired output waveform.

Subsequently, in step 2404, process 2400 includes determining a controlpoint of operation of the power amplifier based on the determinedinstantaneous power level. In an embodiment, the control point ofoperation enhances one or more of linearity and accuracy of the poweramplifier for the determined instantaneous power level. In anembodiment, referring to FIG. 10, step 2404 can be performed by transferfunction module 1006 based on the determined instantaneous power level.

Subsequently, in step 2406, process 2400 includes controlling the poweramplifier to operate according to the determined control point ofoperation. In an embodiment, step 2406 includes performing one or moreof (a) controlling the phase of input signals of the power amplifier;(b) controlling the bias of the power amplifier; and (c) controlling theamplitude of the input signals of the power amplifier. In an embodiment,referring to FIG. 10, step 2406 is performed by transfer function module1006, which accomplishes step 2406 by controlling signals for performing(a), (b), and (c). For example, to control the phase of the inputsignals of the power amplifier, transfer function 1006 may control thesignals it inputs into vector modulators 1008 and 1010. Similarly, tocontrol the bias of the power amplifier, transfer function 1006 may varybias signals 1024 and 1026 that it provides to driver amplifiers 1014and 1016 and MISO amplifier 1018.

According to an embodiment, the control point of operation can be withina first, second, or third control regions, depending on the determinedinstantaneous power level. For example, in an embodiment, the controlpoint of operation is within a first control region when theinstantaneous power level is greater than a first threshold; within asecond control region when the instantaneous power level is greater thana second threshold but lower than the first threshold; and within athird control region when the instantaneous power level is lower thanthe second threshold. According to an embodiment, boundaries of thefirst, second, and third control regions are based on the ComplementaryCumulative Density Function (CCDF) of the desired output waveform.

According to an embodiment of the present invention, when the controlpoint of operation is within the first control region, the controllingstep 2406 of process 2400 includes performing (a) only, or performing(a), (b), and (c). In the latter case, in an embodiment, step 2406includes performing (a) more often than (b) or (c). When the controlpoint of operation is within the second control region, the controllingstep 2406 includes performing (a), (b), and (c). Further, controllingstep 2406 may include performing (b) more often than (a) or (c). Whenthe control point of operation is within the third control region, thecontrolling step 2406 includes performing (b) and (c) only. In anembodiment, controlling step 2406 further includes performing (c) moreoften than (b).

According to an embodiment, controlling step 2406 includes performingone or more of (a), (b), and (c) according to respective weights givento (a), (b), and (c). In an embodiment, the respective weights aredetermined according to one or more of error/system characteristicswithin the power amplifier (e.g., branch phase imbalance, branchamplitude imbalance, branch isolation) and the instantaneous powerlevel.

FIG. 25 illustrates another process flowchart 2500 of a method forcontrol in a power amplifier. Process 2500 begins in step 2502, whichincludes determining a required change in power output from a firstoutput power level to a second output power level in the poweramplifier. In an embodiment, referring to FIG. 10, step 2502 isperformed by transfer function module 1006 based on received I and Qdata reflecting a desired output waveform.

Subsequently, in step 2504, process 2500 includes varying one or moreweights associated with respective power controls of the power amplifierto cause the required change in power output, wherein the power controlsinclude one or more of (a) control of phase of input signals of thepower amplifier, (b) control of bias of the power amplifier, and (c)control of amplitude of the input signals of the power amplifier. In anembodiment, referring to FIG. 10, step 2504 is performed by transferfunction module 1006, which accomplishes step 2504 by varying controlsignals for performing (a), (b), and (c). For example, to control thephase of the input signals of the power amplifier, transfer function1006 may control the signals it inputs into vector modulators 1008 and1010. Similarly, to control the bias of the power amplifier, transferfunction 1006 may vary bias signals 1024 and 1026 that it provides todriver amplifiers 1014 and 1016 and MISO amplifier 1018.

According to an embodiment, the weights associated with the respectivepower controls of the power amplifier are determined according to one ormore of branch phase imbalance, branch amplitude imbalance, and branchisolation within the power amplifier.

According to an embodiment, varying the weights causes the poweramplifier to transition between various classes of operation. Forexample, in an embodiment, varying the weights causes the poweramplifier to transition between class S and class A. In anotherembodiment, varying the weights causes the power amplifier to transitionfrom linear operation to non-linear operation, and vice versa.

FIG. 26 illustrates another process flowchart 2600 of a method forcontrol in a power amplifier. Process 2600 begins in step 2602, whichincludes determining a desired power output trajectory of a desiredoutput waveform of the power amplifier. In an embodiment, referring toFIG. 10, step 2602 can be performed by transfer function module 1006based on received I and Q data reflecting the desired output waveform.

Subsequently, step 2604 includes determining one or more of (a) branchphase imbalance; (b) branch amplitude imbalance; and (c) branchisolation, between branches of the power amplifier. In an embodiment,step 2604 is performed by various error/system measurement modules ofthe power amplifier, which report measurements to transfer functionmodule 1006.

In step 2606, process 2600 includes calculating one or more weightsbased on one or more of the determined branch phase imbalance, branchamplitude imbalance, and branch isolation. In an embodiment, referringto FIG. 10, step 2606 is performed by transfer function module 1006.

Finally, in step 2608, process 2600 includes applying one or more powercontrols according to the one or more weights to control the poweramplifier to generate the desired power output trajectory. In anembodiment, the power controls include one or more of (a) control ofphase of input signals of the power amplifier, (b) control of bias ofthe power amplifier, and (c) control of amplitude of the input signalsof the power amplifier. As noted above, in an embodiment, step 2608 isperformed by transfer function module 1006, which controls differentpower control mechanisms of the power amplifier to apply (a), (b), and(c). For example, to control the phase of the input signals of the poweramplifier, transfer function 1006 may control the signals it inputs intovector modulators 1008 and 1010. Similarly, to control the bias of thepower amplifier, transfer function 1006 may vary bias signals 1024 and1026 that it provides to driver amplifiers 1014 and 1016 and MISOamplifier 1018.

7. CONCLUSION

It is to be appreciated that the Detailed Description section, and notthe Summary and Abstract sections, is intended to be used to interpretthe claims. The Summary and Abstract sections may set forth one or morebut not all exemplary embodiments of the present invention ascontemplated by the inventor(s), and thus, are not intended to limit thepresent invention and the appended claims in any way.

The present invention has been described above with the aid offunctional building blocks illustrating the implementation of specifiedfunctions and relationships thereof. The boundaries of these functionalbuilding blocks have been arbitrarily defined herein for the convenienceof the description. Alternate boundaries can be defined so long as thespecified functions and relationships thereof are appropriatelyperformed.

The foregoing description of the specific embodiments will so fullyreveal the general nature of the invention that others can, by applyingknowledge within the skill of the art, readily modify and/or adapt forvarious applications such specific embodiments, without undueexperimentation, without departing from the general concept of thepresent invention. Therefore, such adaptations and modifications areintended to be within the meaning and range of equivalents of thedisclosed embodiments, based on the teaching and guidance presentedherein. It is to be understood that the phraseology or terminologyherein is for the purpose of description and not of limitation, suchthat the terminology or phraseology of the present specification is tobe interpreted by the skilled artisan in light of the teachings andguidance.

The breadth and scope of the present invention should not be limited byany of the above-described exemplary embodiments, but should be definedonly in accordance with the following claims and their equivalents.

What is claimed is:
 1. A method comprising: partitioning a waveformconstellation space into a plurality of regions, wherein each region ofthe plurality of regions is associated with one or more controlfunctions of a multiple-input-single-output (MISO) device; andtransitioning the MISO device between a plurality of classes ofoperation based on the one or more control functions.
 2. The method ofclaim 1, wherein the partitioning comprises partitioning the waveformconstellation space based on an output power level of a desired outputwaveform from the MISO device.
 3. The method of claim 1, wherein thetransitioning comprises transitioning the MISO device between theplurality of classes of operation based on one or more of a bias controlof the MISO device, a phase control of one or more inputs to the MISOdevice and an amplitude control of the one or more inputs to the MISOdevice.
 4. The method of claim 1, wherein the transitioning comprisesapplying the one or more control functions to the MISO device based on aweighted combination, and wherein the weighted combination associates arespective weight to each of the one or more control functions.
 5. Themethod of claim 4, wherein the applying comprises varying each of therespective weights associated with each of the one or more controlfunctions based on one or more of a branch phase imbalance, a branchamplitude imbalance and a branch isolation of the MISO device.
 6. Themethod of claim 1, wherein the transitioning comprises transitioning theMISO device between classes of operation analogous to Class A, Class B,Class C, Class D and Class S amplifier classes of operation.
 7. Themethod of claim 6, wherein the transitioning comprises operating theMISO device in classes of operation analogous to the Class A and Class Bamplifier classes of operation at a duration of time shorter thanoperating the MISO device in classes of operation analogous to the ClassC, Class D and Class S amplifier classes of operation.
 8. The method ofclaim 1, wherein the transitioning comprises generating a zero, or anear zero, output current with the MISO device when an output waveformof the MISO device undergoes a zero crossing.
 9. A system comprising: amultiple-input-single-output (MISO) device; and a transfer functionmodule configured to transition the MISO device between a plurality ofclasses of operation based on one or more control functions, whereineach the one or more control functions is associated with a region froma plurality of regions partitioned from a waveform constellation space.10. The system of claim 9, further comprising: one or more vectormodulators coupled to respective one or more inputs to the MISO device,wherein the one or more vector modulators are configured to adjust aphase control to one or more input signals received by the MISO device.11. The system of claim 9, further comprising: one or more pre-driverdevices coupled to respective one or more inputs to the MISO device. 12.The system of claim 9, wherein the waveform constellation space ispartitioned based on an output power level of a desired output waveformfrom the MISO device.
 13. The system of claim 9, wherein the transferfunction module is configured to transition the MISO device between theplurality of classes of operation based on one or more of a bias controlof the MISO device, a phase control of one or more inputs to the MISOdevice and an amplitude control of the one or more inputs to the MISOdevice.
 14. The system of claim 9, wherein the transfer function moduleis configured to apply the one or more control functions to the MISOdevice based on a weighted combination, and wherein the weightedcombination associates a respective weight to each of the one or morecontrol functions.
 15. The system of claim 14, wherein the transferfunction module is configured to vary each of the respective weightsassociated with each of the one or more control functions based on oneor more of a branch phase imbalance, a branch amplitude imbalance and abranch isolation of the MISO device.
 16. The system of claim 9, whereinthe transfer function module is configured to transition the MISO devicebetween classes of operation analogous to Class A, Class B, Class C,Class D and Class S amplifier classes of operation.
 17. The system ofclaim 16, wherein the transfer function module is configured to operatethe MISO device in classes of operation analogous to the Class A andClass B amplifier classes of operation at a duration of time shorterthan operating the MISO device in classes of operation analogous to theClass C, Class D and Class S amplifier classes of operation.
 18. Thesystem of claim 9, wherein the transfer function module is configured togenerate a zero, or a near zero, output current with the MISO devicewhen an output waveform of the MISO device undergoes a zero crossing.19. The system of claim 9, wherein the MISO device is configured tooutput a radio frequency (RF) waveform.